1. Field of the Invention
The present invention relates to a receiver in a code division multiple access (to be referred to a CDMA system, hereinafter) system, and more particularly, to a technique for frequency offset estimation used in a spectrum spreading technique.
2. Description of the Related Art
In a code division multiple access (CDMA) system using a spectrum spreading process, data symbols to be transmitted are spread in accordance with a spreading code having a rate higher than a symbol rate. Channels to be multiplexed have different spreading codes and their symbol rates are varied depending on a data rate in transmission. To realize a variable symbol rate without changing a chip rate, a spreading code length per symbol (to be referred to as a spreading rate, hereinafter) shall be controlled. It should be noted that the symbol is a unit for data modulation before the spectrum spreading process is carried out. When the data modulation system is QPSK, one symbol represents a combination of one bit of an in-phase component and one bit of an orthogonal component. That is, the symbol can be expressed by a complex number.
For receiving a spectrum spread signal at high accuracy, it is essential to carry out synchronization detection. For this purpose, it is necessary at a receiver that the frequency of a local signal applied for down-converting an RF (radio frequency) signal to a baseband signal is equivalent to the frequency of a carrier signal from a transmitter. If there is a discrepancy in frequency, i.e., a frequency offset between the local signal at the receiver and the carrier signal at the transmitter, the frequency offset appears on the baseband signal. The frequency offset will cause a timing error in the baseband signal processing or degradation of the S/N ratio after inverse spreading of spectrum, resulting in degrading the quality of a received signal. Particularly, in the CDMA system, the inverse spreading of spectrum of the received signal can not be correctly carried out due to the discrepancy for one chip. Degradation of the S/N ratio after the inverse spreading of spectrum may lead to deterioration of the anti-interference property. Therefore, the development of a higher accuracy automatic frequency controlling system has been desired.
For example, according to a synchronization establishing process of the IMT-2000 technology recommended for international mobile telecommunications, scramble codes on a perch channel are divided to a limited number of groups. For quick acquisition of a cell, the scramble code having a long period is transferred on the channel and a short search code is inserted for every time slot. Orthogonal gold codes are used as the search codes, which are classified into two types, a primary search code and a secondary search code. These search codes are transferred in parallel. The primary search code is a unique code in the system while a plurality of codes are transmitted in a sequence as the secondary code. A mobile terminal receives the primary search code peculiar to the terminal to establish the symbol synchronization and the slot synchronization. In this case, it is desired that the synchronization with the primary search code can be quickly established, and the synchronization with the perch channel can be established. Thus, the cell can be quickly acquired through grouping on the basis of on the scramble code.
FIG. 1 is a block diagram of a conventional automatic frequency controlling apparatus. FIG. 1 shows not the entire structure of a receiver but a relevant section of an automatic frequency controlling process. Also, for simplification of the description, inversely spreading units are limited to two units and such a conventional automatic frequency controlling apparatus is then illustrated.
Referring to FIG. 1, an RF (radio frequency) signal, i.e., a high frequency signal from a transmitter received by an antenna is introduced to a frequency converter 201 via an input terminal 100. The frequency converter 201 receives a first local frequency signal from the first local frequency generator 202. A first local frequency signal is obtained by offsetting the frequency of a carrier signal from the transmitter by an IF frequency. The frequency converter 201 converts the RF signal into an IF (intermediate frequency) signal in accordance with the first local frequency signal. The IF signal is then adjusted to a predetermined signal level by an AGC unit 101 and transferred to an orthogonal demodulator 210. A second local frequency signal having an IF frequency is supplied from a second local frequency generator 203 to an orthogonal demodulator 210. In response to the second local frequency signal, the orthogonal demodulator 210 converts the IF signal into a baseband signal which has a component along an in-phase axis and a component along an orthogonal axis. It is now assumed that QPSK modulation is employed. The in-phase component and the orthogonal component of the orthogonally demodulated signal are passed through two LPF units 102, respectively, and fed to A/D converters 103 which converts into their digital signals. Then, the converted digital signals are transferred to inversely spreading units 220 and a path searching unit 260.
The path searching unit 260 determines a delay profile from the digital signals supplied from the A/D converters 103 to determine the timing for inverse spreading used in the inversely spreading units 220. The intervals for which the delay profile is calculated and the averaged length of the intervals are determined based on an instruction 301 from the controller 300. The path searching unit 260 outputs an inverse spreading timing to the inversely spreading units 220 based on the determined delay profile. Also, the path searching unit 260 determines how many effective multi-paths are present in the received digital signals and delivers its result 303 to the controller 300.
The inversely spreading units 220 receive a control signal 301 from the controller 300. The control signal 301 includes parameter data 301 such as a spreading code and symbol rate of the channel and boundary data of a pilot symbol interval. The inversely spreading units 220 inversely spread the digital signals received from the A/D converters 103 into symbol signals based on the inverse spreading timing received from the path searching unit 260 and the control signal 301. The symbol signals are transferred to pilot symbol inverse demodulators 230. In this conventional example, it is assumed that a pilot symbol signal and a data symbol signal are time-multiplexed in the symbol signal to have a QPSK transmission format, as illustrated in FIG. 2A.
A pilot symbol interval is inserted before a data symbol interval for every slot period having a predetermined interval called “a slot”. A pilot symbol pattern in the pilot symbol interval in each slot period is variable. In this case, the symbol rate can be made variable by changing the spreading rate under a constant chip rate as shown in FIG. 2D. More specifically, the symbol interval in the symbol rate of 2*Fs is decreased to a half of the symbol interval in the symbol rate of Fs, as shown in FIGS. 2B and 2C.
It should be noted that the pilot symbol interval remains unchanged in the length when the symbol rate is varied in FIGS. 2B and 2C. However, there generally is no such a limitation. The pilot symbol interval length may be varied depending on the symbol rate and is not the limitation essential to the present invention.
The controller 300 shown in FIG. 1 receives the number of effective paths 303 from the path searching unit 260. The controller 300 generates a reception channel data such as the spreading code, the symbol rate, and the number of pilot symbols or pilot symbol interval. Also, the controller 300 generates various parameters for frequency offset estimation such as the number of data for phase difference average summation and angle/frequency offset conversion factors. In addition, the controller 300 generates temperature compensated crystal oscillator (TCXO) control data such as a conversion table between frequency offset and TCXO control voltage and the validation or invalidation of an updating operation of frequency offset. The controller 300 supplies the reception channel data as the control signal 301 to the path searching unit 260, the inverse spreading units 220, the pilot symbol inverse modulators 230 and a frequency offset estimator 250. Also, the controller 300 supplies the parameters for frequency offset estimation and a part of the TCXO control data such as the validation or invalidation of the updating operation of frequency offset to the frequency offset estimator 250 as the control signal 301 in addition to the reception channel data. Also, the controller 300 supplies the conversion table between frequency offset and TCXO control voltage to a TCXO controller 270 as the control signal 302.
FIG. 3A is a block diagram of the pilot symbol inverse demodulator 230. In the pilot symbol inverse demodulator 230, a controller 239 generates a generation control signal to the reference pilot symbol generator 232 in response to the control signal 301 from the controller 300. The reference pilot symbol generator 232 generates a pilot symbol pattern for a symbol rate and a concerned slot in response to the generation control signal to output to a pilot symbol inverse demodulator 233. The pilot symbol pattern for the symbol rate and the concerned slot necessary for the inverse demodulation. Thus, the length of the pilot symbol interval is determined based on the control signal 301. The QPSK symbol signal received from the inversely spreading unit 220 is separated by a pilot symbol interval detector 231 into pilot symbols in the pilot symbol interval and data symbols in the data symbol interval based on a control signal form the controller 239. The pilot symbols are delivered to a pilot symbol inverse demodulator 233. The data symbol is subjected to synchronization detection. The pilot symbol inverse demodulator 233 receives the pilot symbol pattern from the reference pilot symbol generator 232 and cancels a modulated component of the pilot symbol signal received from the pilot symbol interval detector 231 to produce an inversely modulated pilot symbol signal. The inversely demodulated pilot symbol signals are then transferred to an addition synthesizer 240 symbol by symbol. The inversely demodulated pilot symbol signals are outputted to the addition synthesizer 240 in the form of a complex vector.
The addition synthesizer 240 complex adds the inversely modulated pilot symbol signals supplied from the two pilot symbol demodulators 230 by a complex adder 251 and outputs the result of the complex addition to the frequency offset estimator 250. The output of the addition synthesizer 240 is expressed as complex vectors.
An example of the inverse demodulation is illustrated in FIGS. 4A and 4B. FIG. 4A shows an example of four pilot symbols received. FIG. 4B illustrates a result of removal or cancellation (or inverse demodulation) of the modulated component of each pilot symbol. When the modulated component of the pilot symbol has been removed, a fluctuation of the transmission path and a frequency offset are obtained at a point after the inverse demodulation.
As shown in FIG. 3B, in the frequency offset estimator 250, a one-symbol delay unit 251 delays the complex vector by one symbol. A complex conjugate multiplier 252 carries out complex conjugate multiplication of a complex vector outputted from the addition synthesizer 240 and the delayed complex vectors outputted from the one-symbol delay unit 251 to calculate a phase difference vector.
Next, based on the control signal 301 from the controller 300, the controller 259 supplies the number of vectors to be averaged and the execution or stop of the averaging operation to the averaging unit 253 and the symbol rate and the execution or stop of the output of the frequency offset expression to the angle/frequency offset converter 255.
The phase difference vectors are then averaged by an averaging unit 253 based on the number of vectors which is designated from a controller 259 which operates based on the control signal 301. It should be noted that the averaging operation by the averaging unit 253 may be a simple summing average, a moving average, or a leak-factor based average. If the path searching unit 260 fails to find an effective path, the averaging operation is not carried out. It is determined based on the designation from the controller 259 which of the averaging operations is carried out or whether the averaging operation is carried out or not.
Next, the phase difference vector averaged by the averaging unit 253 is then converted by an angular converter 254 from the phase difference vector expression to an angular expression. The conversion from the phase difference vector expression to the angular expression can be implemented through arc tangent conversion (arch tan (imaginary part/real part)) using an imaginary part and a real part of the phase difference vector. The angular expression is then transferred to an angle/frequency offset converter 255 where the angular expression is converted to a frequency offset expression in accordance to the symbol rate of the concerned channel designated by the controller 259. The frequency offset expression is transferred to the TCXO controller 270. If no effective path is found by the path searching unit 260, the controller 300 inhibits the updating operation of the frequency offset in the frequency offset estimator 250. Also, if the path searching unit 260 fails to find an effective path, the transfer of the frequency offset expression to the TCXO controller 270 is not carried out.
The TCXO controller 270 has a function to control a voltage applied to the TCXO unit 200 in accordance with the frequency offset value supplied from the frequency offset estimator 250. More particularly, the control voltage applied to the TCXO unit 200 is determined in accordance with the frequency offset using the table designated by the controller 300. In this case, the control voltage applied to the TCXO unit 200 is selected such that the frequency offset is compensated. The control voltage determined by the TCXO controller 270 is a digital value and hence is converted to an analog value by a D/A converter 105 and transmitted via an LPF 102 to the TCXO unit 200.
The first local frequency generator 202 and the second local frequency generator 203 receive a reference local frequency signal from the TCXO 200 with a temperature compensating circuit. The first local frequency generator 202 generates the first local frequency signal which is generated by shifting the frequency of the carrier signal received from the transmitter by the IF frequency. The second local frequency generator 203 generates the second local frequency signal which has the IF frequency.
As described above, in the conventional method, a phase difference vector between symbols is used for estimating the frequency offset. However, the S/N ratio for each symbol is degraded in the transmission frame format in which one slot period is composed of a pilot symbol interval and a data symbol interval as shown in FIG. 2A, as the symbol rate is increased. Hence, there is a problem that the accuracy of estimation of the frequency offset become worse.
More specifically, in the CDMA system in whose frame format a pilot symbol and a data symbol are time multiplexed for transmission, and a variable transmission symbol rate is realized by making the spreading rate variable under a constant chip rate, the spreading rate decreases when the symbol rate increases. As a result, the S/N gain through the spreading process decreases. Accordingly, the frequency offset has to be estimated under a lower S/N ratio condition and its estimation accuracy will be decreased.
In conjunction with the above description, a demodulating method with an adaptable phase control is disclosed in Japanese Laid Open Patent Application (JP-A-Heisei 5-207088). In this reference, a phase control circuit (28) carries out a complex weighting operation to a received complex input signal U such that a square mean of the difference between a desired signal and the complex input signal is made the smallest. A Wiener filter is formed using the phase control circuit (28). A frequency compensating circuit (44) carries out a frequency error estimation based on a variation of a correlation value between the complex input signal U and a demodulation signal D for one symbol period. A phase error estimating circuit (21) carries out an initial phase error estimation based on the frequency error estimation. A phase equalizing circuit (22) carries out a phase equalizing operation in consideration of a phase variation due to the frequency error to fully remove a stationary phase error due to a frequency offset to a correct demodulation signal D.
Also, an accumulation collective demodulator for a K-phase PSK modulated signal is disclosed in Japanese Laid Open Patent Application (JP-A-Heisei 7-202964). In this reference, a complex signal which has been subjected to a quasi-synchronization detection are sampled at a center point iT and a point (i+r)T displaced from the center point to produce (N+1) signals. The (N+1) signals are stored in memories (13 and 24). A estimating section (15) estimates an initial phase error θ'n, and a frequency error Δω' from the signals inputted to the memory (13). Local oscillators (25 and 26) generate local signals exp[−j{θ'0+(Δ'+2kπ/KT)iT}] and exp[−j{θ'0+(Δω'+2kπ/KT) (i+r)T}], respectively. Multipliers (17 and 28) complex multiply the local signals with the signals stored in the memories (13 and 24), respectively. A pattern jitter is removed from the output of the multiplier (28) by a filter (29). An estimating section (27) determines variance of distance from the output of the multiplier (17). The output of the multiplier (17) for k when the variance becomes the least is supplied to the demodulator.
Also, a prediction type synchronization detection apparatus is disclosed in Japanese Laid Open Patent Application (JP-A-Heisei 8-130565). In this reference, reception signals ys(i) which are sampled for every symbol period T are inversely modulated by means (27) into a complex symbol sequence candidates am(i) to am(i−L) (L: is a natural number and L=3 in the figure) to obtain an inverse modulation signal sequence zm(i) to zm(i−L). The inverse modulation signal sequence zm(i−1) to zm(i−L) are weighted and synthesized to produce a front prediction value. Thus, the front prediction error αfm(i) is determined to indicate the difference between the front prediction value and zm (i). The inverse modulation signal sequence zm(i) to zm(i−L+1) are weighted and synthesized to a back prediction value. Thus, a back prediction error αbm(i) is determined to indicate the difference between the back prediction value and zm(i−L). The maximum likelihood estimation is carried out by a maximum likelihood sequence estimating circuit 32 to the summation of squares of each of absolute values of αfm(i) and αbm(i) as the likelihood data and outputs am(i) to am(i−L) and a determination signal. A parameter estimating circuit (47) inputs zm(i) to zm(i−L), αfm(i), αbm(i) and estimates a weight coefficient for producing a prediction value. In this way, characteristic degradation due to a carrier frequency offset and a fading variance can be improved.
Also, a digital mobile radio communication system is disclosed in Japanese Laid Open Patent Application (JP-A-Heisei 9-93302). In this reference, two pilot symbols are provided for one frame. The phase differences between two pilot symbols are added and averaged over a plurality of frames. Thus, a compensation value of a frequency offset is determined to compensate for the frequency offset. In this way, influence due to the frequency offset between a receiver and a transmitter can be reduced in the digital mobile radio system to improve a transmission performance.
Also, a method of receiving a spectrum spread signal and a spectrum spread signal receiving apparatus are disclosed in Japanese Laid Open Patent Application (JP-A-Heisei 11-41141). In this reference, calculation of correlation between a baseband component of a spectrum spread signal and a spreading code is carried out. Then, correlation calculation is carried out at the timing which is different from a timing between the spreading code and the baseband component by ½ of a spreading code interval. The correlation calculation result at the timing which is earlier than ½ of the spreading code interval is estimated using the above calculation results. In this way, a spectrum spread signal receiving apparatus can be made smaller in size and less in power consumption without degradation of the symbol demodulation characteristic, synchronization establishment characteristic, and synchronization tracking characteristic.
Also, a frequency offset correcting apparatus is disclosed in Japanese Patent No. 2,705,613. In this reference, a receiving unit outputs a baseband signal obtained by carrying out demodulation to a reception high frequency signal. An A/D converter converts a baseband signal from the receiving unit into a digital signal. A plurality of correlation processing units carry out inverse spreading to the digital baseband signal from the A/D converter using a spreading signal which is shifted temporally, to produce correlation signals. A plurality of detectors detect the respective correlation signals from the correlation processing units. An addition synthesizer adds synthesizes the detected signals from the detectors. A frequency offset detector compares a signal part of the signal from the addition synthesizer with a theoretical signal of a known signal to detect a frequency offset value. A frequency offset correcting unit removes the frequency offset value detected by the frequency offset detector from the signal outputted from the addition synthesizer for correction.
Also, a data demodulating circuit of a receiving apparatus for a spectrum spreading communication is disclosed in Japanese Patent No. 2,771,757. This reference relates to the data demodulating circuit of the receiving apparatus for the spectrum spreading communication in which a signal which has been subjected to a spectrum spreading operation to an in-phase axis and an orthogonal axis in a direct spreading system is received using a pseudo-noise code in an in-phase axis and a pseudo-noise code in an orthogonal axis and the data is demodulated from the received signal. A receiving signal in the in-phase axis and a receiving signal in the orthogonal axis are multiplied by the pseudo-noise code in an in-phase axis and the pseudo-noise code in an orthogonal axis which correspond to a pilot signal which has been transmitted from a base station, respectively. The multiplication results are integrated. A correlation calculating unit circularly adds and averages the integration result and calculates the correlation which includes a remaining phase difference data after the detection. A phase difference compensating unit compensates for the phase differences which are contained in the received signal in the in-phase axis and the received signal in the orthogonal axis using the phase difference data supplied from the correlation calculating unit.